1. Field of Invention
The present invention relates to a mixer for mixing an AC signal with a reference signal having a particular frequency, and also to a differential amplifier for amplifying the difference between two signals and outputting a resultant amplified differential signal.
2. Description of Related Art
In an RF (Radio Frequency) receiving circuit, a received RF signal is mixed by a mixer with an LO (Local Oscillator) signal and the RF signal is down-converted into an IF (Intermediate Frequency) signal.
FIG. 7 illustrates a mixer in an RF receiving circuit.
FIG. 8 illustrates an exemplary process in which the mixer shown in FIG. 7 down-converts an RF signal into an IF signal.
FIG. 7 illustrates an RF signal serving as a carrier signal and an LO signal supplied from a local oscillator (not shown) applied to the mixer 101. The mixer 101 mixes the RF signal and the LO signal and outputs an IF signal as shown in FIG. 8. Thus, the RF signal is down-converted into the IF signal.
When it is required to remove undesirable signal components in frequency bands other than the IF frequency band from the IF signal obtained via the down conversion, a bandpass filter is generally positioned at a stage following the mixer. FIG. 9 illustrates a mixer and a bandpass filter. FIG. 10 illustrates an exemplary process in which undesirable signal components are removed from an IF signal by the bandpass filter.
As shown in FIG. 10, when there are signals A1 and B1 at both sides of an RF signal, the RF signal including the signals A1 and B1 and the LO signal are applied to the mixer 101 shown in FIG. 9. As a result, in addition to the IF signal, signals A2 and B2 are output from the mixer 101. If signals A2 and B2 are passed through the bandpass filter, the signals A2 and B2 are attenuated into signals A3 and B3. Thus, their influence on the IF signal is reduced.
FIG. 11 illustrates a circuit configuration of the bandpass filter shown in FIG. 9. FIG. 12 illustrates the frequency characteristic of the bandpass filter shown in FIG. 11.
As shown in FIG. 11, the bandpass filter 102 is formed of passive elements including capacitors 102_1 and 102_4 and resistors 102_2 and 102_3. As shown in FIG. 12, the bandpass filter 102 has cutoff frequencies f1 and f2 determined by the values of the passive elements. When the capacitors 102_1 and 102_4 have capacitance C1 and C2, and the resistors 102_2 and 102_3 have resistance R1 and R2, the cutoff frequency f1 is given by the equation:                     f1        =                  1                      2            ⁢                          xe2x80x83                        ⁢            π            ⁢                                          C1                ·                R1                                                                        (        1        )            
and the cutoff frequency f2 is given by the equation:                     f2        =                  1                      2            ⁢                          xe2x80x83                        ⁢            π            ⁢                                          C2                ·                R2                                                                        (        2        )            
The bandpass filter 102 passes frequency components within a particular band determined by the cutoff frequencies f1 and f2.
FIG. 13 illustrates a circuit configuration of a bandpass filter, configured differently from the bandpass filter shown in FIG. 11. FIG. 14 illustrates the frequency characteristic of the bandpass filter shown in FIG. 13.
The bandpass filter 103 shown in FIG. 13 is an active bandpass filter including capacitors 103_1 and 103_4, resistors 103_2 and 103_3, and an operational amplifier 103_5. As with the bandpass filter 102 shown in FIG. 11, the bandpass filter 103 also has cutoff frequencies f3 and f4 determined by values of the passive elements, and the bandpass filter 103 passes frequency components within a particular band determined by the cutoff frequencies f3 and f4.
FIG. 15 illustrates a biquad bandpass filter. FIG. 16 illustrates a circuit configuration of a transconductor amplifier used in the biquad bandpass filter.
The biquad bandpass filter 104 shown in FIG. 15 is a bandpass filter using the Gm-C technology comprising transconductor amplifiers (OTAs: Operational Transconductance Amplifiers) 104_1, 104_2, and 104_3, capacitors 104_4, 104_5, 104_6, and 104_7, and a resistor 104_8. The capacitors 104_4, 104_5, 104_6, and 104_7 all have equal capacitance C, and the resistor 104_8 has resistance R.
The transconductor amplifier 104_1 includes, as shown in FIG. 16, NMOS transistors 104_11, 104_12, 104_13, 104_14, 104_15, 104_16, 104_17, 104_18, and 104_19, constant current sources 104_20, 104_21, 104_22, 104_23, and resistors 104_24 and 104_25. Signals IN+ and INxe2x88x92, which are different in phase by 180xc2x0 from each other, are applied to the NMOS transistors 104_11 and 104_12, respectively. An external voltage signal Vf is applied to the NMOS transistor 14_19. The transconductance gm of the transconductor amplifier 104_1 varies depending on the value of the external voltage signal Vf applied to the NMOS transistors 104_19. The transconductance gm is given by the equation:
gm=xcex2(Vfxe2x88x92Vsxe2x88x92Vt)
wherein xcex2 is the feedback factor of the NMOS transistor 104_19, Vs is equal to Vs2 (when Vs1 greater than Vs2) or Vs1 (when Vs1 less than Vs2) (Vs1 and Vs2 are source and drain voltages, respectively, of the NMOS transistor 104_19), and Vt is the threshold voltage of the NMOS transistor 104_19.
Although the circuit configuration has been described above only for the transconductor amplifier 104_1, the transconductor amplifiers 104_2 and 104_3 also have a similar circuit configuration.
FIG. 17 illustrates the frequency characteristic of the biquad bandpass filter shown in FIG. 15.
The frequency characteristic of this biquad bandpass filter 15 shown in FIG. 17 is variable. More specifically, the cutoff frequencies f01 and f02 can be varied by varying the external voltage signal Vf thereby varying the transconductance gm of the transconductor amplifiers 104_1, 104_2, and 104_3. For example, when the external voltage signal Vf applied to the transconductor amplifier 104_2 is varied, the center frequency f0 shown in FIG. 17 is given by the equation:
f0=gm2/2xcfx80C
where gm2 is the transconductance of the transconductor amplifier 104_2.
On the other hand, the difference between the cutoff frequency f01 and the cutoff frequency f02 is given by the equation:
xcex94f=gm2xc3x97R.
In the bandpass filters 102 and 103 shown in FIGS. 11 and 13, respectively, their cutoff frequencies are determined by the values of passive elements. This means that, to change the cutoff frequencies, the passive elements themselves must be changed. To change the values of passive elements formed on a semiconductor chip using CMOS technology or the like, it is required to change the layout of the passive elements of the semiconductor chip. The change in the layout needs a long time and high cost and thus the change results in great disadvantages in production or development. Another problem is that passive elements occupy large areas on the semiconductor chip.
Although the biquad bandpass filter 104 shown in FIG. 15 has the advantage that the cutoff frequencies can be controlled by the external voltage signal, the biquad bandpass filter 104 has the disadvantage that the circuit configuration of the transconductor amplifiers 104_1, 104_2, and 104_3 is complicated, needs a large number of transistors, and then needs a large-scale circuit.
In view of the above, it is an object of the present invention to provide a mixer and a differential amplifier which have simple circuit configurations and which allow the cutoff frequency to be easily changed.
According to an aspect of the present invention, a mixer is provided for mixing an AC signal with a reference signal having a particular frequency, wherein the mixer includes a parallel resonant circuit including an active inductor and serving as an output load.
Preferably, the AC signal is an RF signal and the reference signal is an output signal of a local oscillator, the frequency of the output signal being different by a particular value from the frequency of the RF signal.
Preferably, the active inductor includes two transconductance circuits and a capacitor such that the inductance of the active inductor is set by the transconductance of the two transconductance circuits and the capacitance of the capacitor.
Preferably, the inductance of the active inductor can be arbitrarily varied by controlling the transconductance of the two transconductance circuits in response to an external signal.
Preferably, the parallel resonant circuit comprises an active inductor, a capacitor, and a resistor that are connected in parallel.
Preferably, the parallel resonant circuit has bandpass frequency selectivity given by the expression:             1              2        ⁢                  xe2x80x83                ⁢        π        ⁢                  LC                      +                  R        2            ·                        C          L                      ≥  f  ≥            1              2        ⁢                  xe2x80x83                ⁢        π        ⁢                  LC                      -                  R        2            ·                        C          L                    
where L is the inductance of the active inductor, C is the capacitance of the capacitor, R is the resistance of the resistor, and f is the resonant frequency of the parallel resonant circuit.
According to another aspect of the present invention, there is provided a differential amplifier for amplifying the difference between two input signals and outputting a resultant amplified differential signal, wherein the differential amplifier includes a parallel resonant circuit including an active inductor and serving as an output load.
Preferably, the active inductor includes two transconductance circuits and a capacitor such that the inductance of the active inductor is set by the transconductance of the two transconductance circuits and the capacitance of the capacitor.
Preferably, the inductance of the active inductor can be arbitrarily varied by controlling the transconductance of the two transconductance circuits in response to an external signal.
Preferably, the parallel resonant circuit includes an active inductor, a capacitor, and a resistor that are connected in parallel.
Preferably, the parallel resonant circuit has bandpass frequency selectivity given by the expression:             1              2        ⁢                  xe2x80x83                ⁢        π        ⁢                  LC                      +                  R        2            ·                        C          L                      ≥  f  ≥            1              2        ⁢                  xe2x80x83                ⁢        π        ⁢                  LC                      -                  R        2            ·                        C          L                    
where L is the inductance of the active inductor, C is the capacitance of the capacitor, R is the resistance of the resistor, and f is the resonant frequency of the parallel resonant circuit.
As described above, each of the mixer and the differential amplifier according to the present invention has a resonant circuit including an active inductor and serving as an output load. The active inductor includes a transconductance circuit, which is constructed in a simple form as will be described later with reference to specific embodiments. The inductance L of the active inductor can be arbitrarily varied by controlling the transconductance of the transconductance circuit in response to an external signal, thereby easily varying the cutoff frequencies of the bandpass frequency selectivity. Each of the mixer and the differential amplifier can be formed on a semiconductor chip with a smaller size than is needed for a conventional bandpass filter using passive elements.